Communication systems are frequently divided into wireless communication systems such as, e.g., mobile radio systems and wireless networks, and wire-connected communication systems. The Universal Mobile Telecommunications System (UMTS) is a current example of a mobile radio system. The basic architecture of a UMTS mobile radio system exhibits, among other things, mobile stations (User Equipment (UE)) and a radio access network (UMTS Terrestrial Radio Access Network (UTRAN)). The radio access network contains facilities for transmitting data via radio, such as, e.g., base stations which are called Node B in UMTS mobile radio systems. The base stations in each case supply a particular area and a cell, respectively, in which mobile stations can be located. The interface between a mobile station and a base station which communicate wirelessly by radio is called a radio interface (Uu Interface).
In the description following, reference is made to specifications 3GPP TS 25.213, V5.3.0 (2003-03), Spreading and modulation (FDD), 3GPP TS 25.104, V6.2.0 (2003-06), Base Station (BS) radio transmission and reception (FDD) and 3GPP TS 25.141, V6.2.0 (2003-06), Base Station (BS) conformance testing (FDD) of the 3rd Generation Partnership Project (3GPP), Technical Specification Group Radio Access Network.
In a UMTS mobile radio system, digital data to be transmitted on the route from a base station to a mobile station are initially subjected to channel coding. This provides the digital data with redundancy and protects them against faulty transmission via a disturbed mobile radio channel or, respectively, enables an error correction to be performed at the respective receiver of the data. The digital data are then divided into two data bit streams by a serial/parallel conversion and mapped onto symbols in a complex plane by means of a digital modulation method. The digital data are then distributed to physical channels within the capacity of the available transmission bandwidth by means of a multiple access method. The physical channels are combined with one another and then scrambled, for example station-specifically, by means of a scrambling code. To transmit the digital data via a mobile radio channel, orthogonal carrier signals are used. The mobile radio channel is divided for transmitting operation and receiving operation by means of a duplex method.
The digital modulation method used in the UMTS mobile radio system is quaternary phase shift keying (QPSK) or 16-quadrature amplitude modulation (16QAM), in which in each case two (QPSK) or four (16QAM) successive bits of a bit sequence to be transmitted are combined and are in each case mapped to a symbol of a symbol space spanned by a real in-phase branch (I) and an imaginary quadrature branch (Q) in the complex plane, which has four (QPSK) or 16 (16QAM) elements.
In the UMTS standard or in the 3GPP standard, respectively, the code division multiple access method (CDMA) is used as multiple access method in which a bipolar data bit stream to be transmitted is multiplied by a subscriber-specific bipolar code sequence or a spreading code and is spread. The elements of the spreading code are called chips in order to be able to semantically distinguish them from the bits of the data bit stream. In principle, chips are nothing other than bits. The multiplication of the data bit stream by the chip stream results again in a bipolar data stream. In general, the rate of the chip stream is a multiple of the rate of the data bit stream and this is determined by the length of the spreading code which is specified by a spreading factor (SF). The spreading factor corresponds to the number of chips per bit. With a constant chip rate on the radio transmission link between transmitters and receivers, the data bit rate represented in the chip stream is only dependent on the spreading factor of the respective subscriber-specific spreading code. In the UMTS mobile radio system, orthogonal spreading codes with variable spreading factor (OVSF=Orthogonal Variable Spreading Factor) are used in order to be able to implement variable data rates. The OVSF spreading codes are defined in specification 3GPP TS 25.213. In this arrangement, the data rate can fluctuate within a range from 32 kbit/s to 2 Mbit/s.
The wideband code division multiple access method (WCDMA) has been chosen by the ETSI (European Telecommunications Standard Institute) as the basis for the FDD UMTS radio interface (Uu Interface) in which an operation at the same data rate in both directions of transmission or, respectively, symmetric uplink/downlink operation is possible. According to the UMTS standard, data are transmitted between the base stations and the mobile stations in frames. Each frame has in each case 15 slots which in each case contain 2560 chips. One frame has a duration of 10 ms and a slot thus has a duration of 666 μs and a chip has a duration Tc of about 0.2604 μs. The chip rate Fchip is 38 400 chips per frame or, respectively, 3.84 Mchips/s.
To separate transmit signals from receive signals of a base station or a mobile station or, respectively, separate the uplink from the mobile station to the base station from the downlink from the base station to the mobile station, the time division duplex method (TDD) or the frequency division duplex method (FDD) can be used in UMTS mobile radio systems. In the FDD method, the stations transmit and receive in each case separate frequency bands. In this arrangement, the transmit band of one station is the receive band of the other station and vice versa.
All subscribers use the multiple access method for impressing a fingerprint on their useful data by means of a subscriber-specific spreading code which allows the transmitted signal to be restored from the sum of the received signals. Various data bit streams which are to be transmitted in parallel from a transmitter are multiplied by orthogonal spreading codes in the real in-phase branch and the imaginary quadrature branch of the modulation method and are then added. The complex sum signal is then also scrambled with a specific complex scrambling code by means of a chip-wise and frame-aligned complex multiplication of the sum signal. In the FDD mode of the UMTS mobile radio system, the scrambling code is station-specific, i.e. each base station and each mobile station can use a different scrambling code. From the received chip sequence, the bits of the data bit stream can be recovered in a receiver by repeating the process of multiplication. For this purpose, the chip stream is again multiplied or correlated in the correct phase with the same spreading code which was already used in the transmitter which again results in the transmitted data bit stream.
On the uplink, for example, information is transmitted from the mobile stations via a radio link to the base stations. The information of various mobile stations is coded in accordance with the CDMA multiple access method and transmitted in physical channels which are combined to form one radio signal via a common frequency channel or radio channel to the base stations with which the mobile stations are in radio contact. In the FDD mode, a physical channel is defined by the spreading code, which is also called a channelization code, and by the frequency channel.
Generally, a distinction is made between so-called dedicated physical channels and common physical channels. A dedicated physical channel is exclusively used by one connection and is reallocated during the call set-up and possibly during the call. Common physical channels are simultaneously or alternately utilized by a number of calls.
In the FDD mode, physical channels are, for example, the dedicated physical channel (DPCH), the common control physical channel (CCPCH), the common pilot channel (CPICH) and the synchronization channel (SCH). The SCH is a channel of the downlink and is used for cell search and synchronization of the mobile stations. It is subdivided into two subchannels, the primary synchronization channel (P-SCH) and the secondary synchronization channel (S-SCH). The P-SCH is identical for all cells and is, therefore, sent out without scrambling.
FIG. 12 shows the arrangement of a transmitter or, respectively, a modulation arrangement which is defined in specification 3GPP TS 25.213 for the downlink, i.e. from a base station to a mobile station. In this case, the transmitter uses a particular carrier frequency of ω/2π. A bit stream of a plurality of bit streams with different bit rates is in each case converted by a serial-parallel converter (S->P) into in each case two parallel bit streams. The two bit streams are mapped onto symbols in the complex plane by a mapping device according to a digital modulation method such as, e.g., QPSK or 16QAM (Quadrature Amplitude Modulation). The symbol streams allocated to the in-phase branch (I) and to the quadrature branch (Q) are spread with a respective spreading code Cch,SF,m in multiplication devices. In this arrangement, a different spreading code is allocated to each physical channel. The resultant real chip streams of the quadrature branch (Q) and of the in-phase branch (I) are converted by addition into a single complex chip stream and this chip stream is then scrambled or encoded by means of a complex scrambling code Sdl,n. After the scrambling, each physical channel is weighted (point S) by means of a weighting factor Gi in accordance with its power level. The resultant chip streams are combined in an addition device and in a further addition device, the synchronization channels P-SCH and S-SCH weighted with the weighting factors GP and GS are added to the complex chip stream. The resultant chip stream is divided into a real component and an imaginary component in a device (point T). The spectra of the real component and of the imaginary component are shaped in two shape filters such as, e.g., RRC (square Root Raised Cosine) filters, which exhibit a roll-off of 20%. The filtered chip stream or, respectively, the signal representing it is finally up-converted by mixers to the particular carrier frequency using orthogonal carrier signals, and conducted to a power amplifier (not shown) and to a transmitting antenna (not shown).
The signal which is theoretically obtained at the transmitting antenna is a reference signal. The definition of the WCDMA downlink signal is specified in specification 3GPP TS 25.213. To facilitate the implementation of a transmitter, the UMTS standard according to specifications 3GPP TS 25.104 and 3GPP TS 25.141 allows an actual signal to differ slightly from the reference signal and defines quality requirements which must be met. The quality requirements are defined by reference signals which must be used during tests (test modes 1 to 5) and quality factors which must be measured during tests. These are the error vector magnitude (EVM), the peak code domain error (PCDE) and the adjacent channel leakage power ratio (ACLR).
The chip stream or, respectively, the signal at point T is a sum of many complex random signals of various physical channels. The result of this sum is a Gaussian signal which disadvantageously has a wide dynamic range. The filtering operations following point T can further increase the dynamic range of the signal if the spreading codes have not been suitably selected.
FIG. 13 shows the complementary cumulative distribution function (CCDF) of a WCDMA signal, both at point T and at the output of the transmitter of FIG. 12. The vertical axis represents the probability of the instantaneous power of a signal being greater than the value along the horizontal axis. The probability value used for defining the dynamic range of the signal is 10−4. FIG. 13 shows that the power of the WCDMA signal at point T exhibits a dynamic range of 10 dB with respect to the root mean square (RMS) of the power. Furthermore, it can be seen from the complementary cumulative distribution function for the output of the transmitter that the dynamic range becomes greater if the processing following point T in the transmitter is taken into consideration.
The device in the transmitter influenced most by a wide dynamic range of the generated signal is the power amplifier which, among other things, is the most complex and expensive item. The power amplifier must have a linear characteristic over the entire input range. With a WCDMA signal, the operating point of the power amplifier must be far away (10 dB or more) from its saturation point in order to avoid nonlinear effects and signal leakage into adjacent channels. The result of this is that the power amplifier must be selected to be oversized with respect to the desired rms power; that a larger cooling system is required for the larger power amplifier since the efficiency of a power amplifier is very low; that the requirements for setting the operating point of the power amplifier are higher; and that the power consumption of the power amplifier is greater. In summary, this is associated with greater expenditure and with higher costs for the operator of a mobile radio system.
WO 99/53625 A1 describes an implementation of an amplitude limiting system in a CDMA system which is described in the text which follows.
FIG. 14A firstly shows an implementation of a conventional transmitter. As in FIG. 12, a symbol source generates complex symbols with a symbol rate of 1/T. When the symbol source generates a WCDMA signal, its symbol rate or chip rate is 3.84 MHz. The complex symbols are fed into a shape filter which generates at an output of said filter an interpolated version of the symbol sequence with a symbol rate increased by K and a desired spectrum. The symbol sequence is then mixed to an intermediate frequency in a mixer and converted into a continuous analogue time domain signal using a digital/analogue converter (DAC). After further analogue processing in a processing device which exhibits several stages of band-pass filtering and mixing to the suitable carrier frequency, the signal is conducted to a power amplifier and finally to a transmitting antenna.
FIG. 14B shows the transmitter of FIG. 14A with a precompensation device between the symbol source and the shape filter as described in WO 99/53625 A1. The precompensation device is used for reducing the dynamic range of a signal to be transmitted. FIG. 15 also shows a diagrammatic representation of the precompensation device. The amplitude of the symbols of a sequence x(nT) of an input signal is compared with a threshold value in the precompensation device. When this threshold value is exceeded, a damping factor a(nT) is calculated. From the multiplication of the symbols with the damping factor x(nT)*a(nT) in a multiplication device, a sequence xc(nT) of a corrected signal or, respectively, of an output signal of the precompensation device is obtained. The damping factor a(nT) is determined as follows:if |x(nT)|>threshold valuea(nT)=1−(1−threshold value/|x(n)|)elsea(nT)=1  (1)
A significant advantage of the transmitter in FIG. 14B consists in that due to the fact that the correction is carried out before the shape filter, no leakage from the desired signal band can be expected. However, the transmitter has a significant disadvantage which will be illustrated by means of an example. Let it be assumed that the symbol source generates one of the following three sequences of symbols:1) +1 0 0 0 +1 0 0 0 +1 0 0 0 +1 02) +1 +1 +1 +1 0 0 0 0 0 0 0 0 0 03) −1 +1 +1 −1 0 0 0 0 0 0 0 0 0 0  (2)
These three sequences have the same energy and the same dynamic range. The sequences are then sent to the shape filter which is, e.g., an RRC filter. At the output of the shape filter, the three sequences generate signals with very different dynamic ranges. The third sequence, in particular, generates, at the output of the shape filter, a sequence with a peak which is greater than a peak in a sequence which is generated at the output of the shape filter by the first sequence. Since the shape of the signal, after passing through a number of interpolation stages in the transmitter, is unknown for the precompensation device of FIG. 15, the threshold value must be set to a very low value in the precompensation device in order to achieve the desired dynamic range in the power amplifier. This influences all sequences in the same manner and the system does not distinguish between sequences in which no correction is required and sequences in which a correction is required. The result of this is that unnecessary disturbances or distortions are introduced into a signal or a sequence.
Therefore, a significant disadvantage of the transmitter of FIG. 14B and of the precompensation device of FIG. 15, respectively, consists in that the precompensation device cannot distinguish when a correction of a sequence of a signal is required and when not, and, therefore, cannot effect an efficient reduction of the dynamic range of a signal in a transmitter without unnecessarily distorting the signal.
A further known possibility for reducing the dynamic range of a signal consists in using FIR filters for generating a correction signal from a sequence of an amplitude correction signal, the correction signal being added to the signal.
In T. May, H. Rohling, “Reducing the Peak-to-Average Power Ratio in OFDM Radio Transmission Systems”, Proc. IEEE VTC '98, Phoenix, May 1998, pp. 2472-2478 and N. Hentati, M. Schrader, “Additive Algorithm for Reduction of Crest Factor”, 5th International OFDM Workshop 2000, Hamburg, pp. 27-1-27-5, devices and methods for reducing the dynamic range of orthogonal frequency division multiplex (OFDM) signals are described.
Both T. May et al. and N. Hentati et al. achieve the reduction of a crest factor of a signal by adding a correction signal to the signal.xc(n)=x(n)+c(n)  (3)x(n) is a sampling sequence or sequence of a signal to be corrected, c(n) is a sequence of a correction signal and xc(n) is a sequence of a corrected signal. The sequence of the correction signal is obtained as follows:
                              c          ⁡                      (            n            )                          =                              ∑                          k              =                              -                ∞                                      ∞                    ⁢                                                    h                                  n                  -                  k                                            ·              Δ                        ⁢                                                  ⁢                          c              ⁡                              (                k                )                                                                        (        4        )            h(.) is a pulse which is used for the correction, and Δc(k) is a sequence Δc(k) of an amplitude correction signal which is used for reducing the crest factor to a target value. In T. May et al. and N. Hentati et al., the sequence Δc(k) of the amplitude correction signal is obtained by using the following expression:Δc(k)=x(k)−x(k)/|x(k)|·threshold value  (5)
The pulse h(.) is either a Gaussian or an si-function-like (si=sinx/x) pulse. An impulse response of a complex band-pass filter can be used for signals which are allocated to frequencies not equal to zero. One possibility for generating the pulses h(.) is using a filter with a finite impulse response (FIR). In the case where the pulse or, in this case, the impulse response of the FIR filter has a length of 2N+1 coefficients, equation 4 receives the form
                              c          ⁡                      (            n            )                          =                              ∑                          k              =                              -                N                                      N                    ⁢                                                    h                                  n                  -                  k                                            ·              Δ                        ⁢                                                  ⁢                          c              ⁡                              (                k                )                                                                        (        6        )            
FIG. 16 shows a device for reducing the dynamic range of an OFDM signal. The device exhibits an amplitude correction signal generating device, an FIR filter with delay elements (z−1) and coefficients hi, a delay device and an addition device. The coefficients hi of the FIR filter are 2N+1 samples of a pulse h(n). In T. May et al., it is assumed that the main coefficient h0 is equal to 1. The amplitude correction signal generating device then generates a sequence Δc(k) of an amplitude correction signal according to equation 5. If the sequence Δc(k) of the amplitude correction signal differs from zero, the output signal of the FIR filter is equal to the sequence c(n) of the desired correction signal after N+1 steps. If the delay of the delay device for the sequence x(k) of the signal to be corrected is equal to N+1, the sequence c(n) of the correction signal of the FIR filter is aligned with the sequence x(n) of the delayed signal to be corrected. If the sequence c(n) of the correction signal is subtracted from the sequence x(n) of the delayed signal in the addition device, a sequence xc(n) of the corrected signal having a desired dynamic range is generated.
FIG. 17 shows the sequence c(n) of the correction signal which is generated by the FIR filter, and a sequence Δc(k) of the associated amplitude correction signal which is shown as a narrower curve with points for a case in which the target dynamic range or, respectively, the threshold value is large. The filtered sequence of the amplitude correction signal or, respectively, the sequence c(n) of the correction signal exactly matches the sequence Δc(k) of the amplitude correction signal fed into the FIR filter at the correction times. The sequence x(n) of the signal to be corrected exceeds the desired threshold value with a very low probability and corrections are only necessary occasionally.
FIG. 18 shows the sequence c(n) of the correction signal for a case in which the target dynamic range or, respectively, the threshold value is of medium magnitude. The corrections already occur with such frequency that the pulses of the sequence c(n) of the correction signal overlap. Nevertheless, the correspondence between the sequence c(n) of the correction signal and the sequence Δc(k) of the amplitude correction signal at the input of the FIR filter is good at the correction times. The correction signal of the FIR filter is slightly too large for the middle correction value.
FIG. 19 shows the sequence c(n) of the correction signal which is generated by the FIR filter for the case in which the threshold value of the dynamic range is set to a very low value and, therefore, the pulses of the sequence of the correction signal overlap much more. The correspondence between the sequence c(n) of the correction signal and the values of the sequence Δc(k) of the amplitude correction signal at the input of the FIR filter is poor at the correction times.
A disadvantage of the device for reducing the dynamic range of an OFDM signal in FIG. 16, therefore, consists in that with a very low threshold value of the dynamic range, overcorrection of the signal to be corrected occurs and the overcorrection introduces unnecessary noise into the signal to be corrected. The problem of overcorrection is based on the fact that the tails of old pulses of the correction signal overlap new pulses.
A further disadvantage of the device consists in that possibly unwanted larger peaks are generated outside the correction times.
In some countries, the operators of a mobile radio system can use more than one carrier frequency in a cell of a mobile radio system. These carrier frequencies are located in adjacent frequency channels with a spacing of, for example, about 5 MHz. In such mobile radio systems, therefore, it is necessary to reduce the dynamic range of signals to be transmitted for all carrier frequencies.
FIG. 20 shows a conventional transmitter for generating a number of carrier signals (N) having different carrier frequencies. To implement such a transmitter, a basic transmitter structure is repeated for the N carriers. The basic transmitter structures in each case exhibit mapping devices for digital modulation such as, e.g., QAM modulation, of bit streams bi, weighting devices for weighting the bit streams with weighting factors Gi, a spreading code generator for spreading the weighted bit streams with spreading codes CCH,SF,m and for generating chip streams (physical channels) with a chip rate Fchip, a first addition device for combining the physical channels to form one signal, a scrambling code generator for scrambling the signal (point S), a second addition device for adding synchronization channels to the signal (point T), a shape filter, a processing device and a power amplifier, for various physical channels. The N carrier signals of the N basic transmitter structures are added in a third addition device and the resultant multi-carrier signal is conducted to an antenna.
FIG. 21 shows a further conventional transmitter for generating a number of carrier signals having different carrier frequencies. This transmitter is similar to the transmitter of FIG. 20 but can be implemented with much lower expenditure. In the transmitter of FIG. 21, each carrier signal of a basic transmitter structure is mixed to an intermediate frequency f by means of a mixer following the shape filter. The frequency spacing between the carrier signals already corresponds to the required frequency spacing. The carrier signals of the individual basic transmitter structures are then added outside these by means of the third addition device in order to generate a multi-carrier signal. The multi-carrier signal is then finally up-converted to a centre radio frequency in a processing device and sent to a single multi-carrier power amplifier and to an antenna. Since the power amplifiers are the most elaborate components in the transmitters, the reduction in complexity of the transmitter of FIG. 21 in comparison with the transmitter of FIG. 20 becomes clear immediately. In the transmitter of FIG. 20, a power amplifier is provided in each basic transmitter structure whereas only one multi-carrier power amplifier is needed in the transmitter of FIG. 21.
However, a disadvantage of the transmitter in FIG. 21 consists in that the multi-carrier signal has a dynamic range which is greater than the dynamic range of the individual carrier signals as is also shown in FIG. 13. This makes it more difficult to meet the requirements for the multi-carrier power amplifier and it is necessary to reduce the wide dynamic range.
In multi-carrier WCDMA Basestation Design Considerations —Amplifier Linearization and Crest Factor Control—Technology White Paper—PMC Sierra Inc.—Issue 1: Aug. 1, 2002, Andrew Right and Oliver Nesper describe devices for reducing the dynamic range of a multi-carrier signal.
FIG. 22 shows such a device for reducing the dynamic range of a multi-carrier signal which can be used for a transmitter in FIG. 21 from point T onward. N signals which in each case have a chip rate Fchip are firstly interpolated to a higher rate by means of a shape filter such as, e.g., an RRC filter or an equivalent device. At this rate, it is possible to mix the respective signal to an intermediate frequency f1, f2, . . . , fN by means of an associated mixer. The N generated carrier signals x1, x2, . . . xN at outputs of the mixers have the desired spacing between the carrier frequencies. At this point, the carrier signals are added by means of a first addition device in order to generate a reference multi-carrier signal v. If the amplitude/power of the reference multi-carrier signal v exceeds a threshold value which is fixed or variable, a multi-carrier amplitude correction signal Δv is calculated by means of a multi-carrier amplitude correction signal generating device in order to push the reference multi-carrier signal v below the threshold value. The reference multi-carrier signal v, the multi-carrier amplitude correction signal Δv and the carrier signals x1, x2, . . . xN are then fed into a correction pulse generator which generates from these N complex single-carrier correction signals c1, c2, . . . cN. The single-carrier correction signals c1, c2, . . . cN are in each case pulses h1(.), h2(.), . . . , hN(.) modulated with single-carrier amplitude correction signals Δc1, Δc2, . . . ΔcN. Each single-carrier correction signal c1, c2, . . . cN is in each case allocated to a carrier signal x1, x2, . . . xN. The single-carrier correction signals c1, c2, . . . cN are finally added to a delayed version of each carrier signal x1, x2, . . . xN in second addition devices in order to generate corrected carrier signals. The delayed carrier signals are generated by delay devices and the respective delay equalizes the processing time of the first addition device, the multi-carrier amplitude correction signal generating device and the correction pulse generator and the group delay of the pulses. The N corrected carrier signals are then added together in a third addition device to form a multi-carrier signal and are sent to further up-conversion stages and, finally, to the multi-carrier power amplifier as shown in FIG. 21. The pulses hi(.) have the function of reducing the spectral power density of an error within the carrier band in such a manner that the requirements for the adjacent-channel leakage ratio (ACLR) are met. The N pulses hi , which are modulated with the single-carrier amplitude correction signals Δc1, Δc2, . . . ΔcN, when added together, form a multi-carrier correction signal. If the multi-carrier correction signal is added to a properly delayed multi-carrier signal, its amplitude at the sampling position t0 at which originally a peak was detected is reduced. These pulses can be low-pass pulses which are then up-converted to the relative carrier frequency, or complex band-pass pulses which are designed for the carrier band.
FIG. 23 shows a diagram which represents the effect of the device of FIG. 22, considering N=4 carrier signals. The four carrier signals are represented by the vectors x1, x2, x3 and x4 in the complex plane. When the carrier signals are added, they generate the reference multi-carrier signal v or, respectively, the vector of the reference multi-carrier signal v. The vector of the reference multi-carrier signal v is outside a target region which is defined by the threshold value. The target region is represented by complex points within a circle. The device of FIG. 22 generates a multi-carrier amplitude correction signal Δv which reduces the amplitude (power) of the reference multi-carrier signal v in the target region.
The multi-carrier amplitude correction signal is obtained from Δv=−α·v and is distributed over the four carrier signals xi. FIG. 23 shows in this regard the choice of the N single-carrier amplitude correction signals known from Andrew Right et al.:Δc1=−α·x1, Δc2=−α·x2, . . . , ΔcN=−α·xN  (7)
FIG. 23 shows that the vectorial addition of the four single-carrier amplitude correction signals Δci corresponds to the multi-carrier amplitude correction signal Δv. At time (t0), at which the amplitude reduction is necessary, the single-carrier amplitude correction signals Δci reduce the amplitude in the target region. For different times, the multi-carrier amplitude correction signal Δv is the sum of the pulses(Δc1·h1(t−t0), Δc2·h2(t−t0), . . . ΔcN·hN(t−t0))  (8)t0 is the time at which the amplitude correction is necessary.
A disadvantage of selecting the single-carrier amplitude correction signals according to equation 7 consists in that they cause too great a distortion of the multi-carrier signal. The power of this distortion is proportional to the energy of the pulses used and to the amplitude of the single-carrier amplitude correction signals used. The pulses possibly have the same energy.
FIG. 24 shows a diagram which represents the effect of the device of FIG. 22 for the case in which a vector, in this case the vector of the carrier signal x3, is not the cause of the large amplitude of the reference multi-carrier signal v. In the calculation of the single-carrier amplitude correction signals according to equation 7, the single-carrier amplitude correction signal Δc3=−α·x3 actually represents too much correction, which introduces unnecessary noise into the carrier signal x3.
In the cases described above, signals are considered which themselves are a sum of a number of signals. Each of these signals has different frequency bands and can be generated by different modulation methods. These can be individual carrier signals or a number of carrier signals which are modulated with different modulation methods such as, e.g., the quaternary amplitude modulation (QAM) or the quaternary phase shift keying (QPSK) and are coded with multiple access methods (CDMA, WCDMA etc.) or coded multi-carrier techniques (OFDM=Orthogonal Frequency Division Multiplex). These signals have a dynamic range which fluctuates greatly and is in some cases wide and which leads to great complexity in the circuits, and particularly amplifiers, used in a transmitter.